Printed circuit board employing lossy power distribution network to reduce power plane resonances

ABSTRACT

An interconnecting apparatus employing a lossy power distribution network to reduce power plane resonances. In one embodiment, a printed circuit board includes a lossy power distribution network formed by a pair of parallel planar conductors separated by a dielectric layer.

BACKGROUND OF THE INVENTION

[0001] 1. Field of the Invention

[0002] This invention relates to electronic systems, and moreparticularly to power distribution networks embodied within printedcircuit boards and semiconductor device packages having continuousplanar conductors.

[0003] 2. Description of the Related Art

[0004] Electronic systems typically employ several different types ofelectrical interconnecting apparatus having planar layers ofelectrically conductive material (i.e., planar conductors) separated bydielectric layers. A portion of the conductive layers may be patternedto form electrically conductive signal lines or “traces”. Conductivetraces in different layers (i.e., on different levels) are typicallyconnected using contact structures formed in openings in the dielectriclayers (i.e., vias). For example, printed circuit boards typically haveseveral layers of conductive traces separated by dielectric layers. Theconductive traces are used to electrically interconnect terminals ofelectronic devices mounted upon the PCB. Similarly, semiconductor devicepackages often have several layers of conductive traces separated bydielectric layers to electronically connect bonding pads of anintegrated circuit to terminals (e.g., pins or leads) of the devicepackage.

[0005] Signals in digital electronic systems typically carry informationby alternating between two voltage levels (i.e., a low voltage level anda high voltage level). A digital signal cannot transitioninstantaneously from the low voltage level to the high voltage level, orvice versa. The finite amount of time during which a digital signaltransitions from the low voltage level to the high voltage level iscalled the rise time of the signal. Similarly, the finite amount of timeduring which a digital signal transitions from the high voltage level tothe low voltage level is called the fall time of the signal.

[0006] Digital electronic systems are continually being produced whichoperate at higher signal frequencies (i.e., higher speeds). In order forthe digital signals within such systems to remain stable for appreciableperiods of time between transitions, the rise and fall times of thesignals must decrease as signal frequencies increase. This decrease insignal transition times (i.e., rise and fall times) creates severalproblems within digital electronic systems, including signal degradationdue to reflections, power supply “droop”, ground “bounce”, and increasedelectromagnetic emissions.

[0007] A signal driven upon (i.e., launched) from a source end of aconductive trace suffers degradation when a portion of the signalreflected from a load end of the trace arrives at the source end afterthe transition is complete (i.e., after the rise time or fall time ofthe signal). A portion of the signal is reflected back from the load endof the trace when the input impedance of the load does not match thecharacteristic impedance of the trace. When the length of a conductivetrace is greater than the signal transition time (i.e., the rise or falltime) divided by about 20 times the delay per unit length along thetrace, the effects of reflections upon signal integrity (i.e.,transmission line effects) should be considered. If necessary, stepsshould be taken to minimize the degradations of signals conveyed uponthe trace due to reflections. The act of altering impedances at thesource or load ends of the trace in order to reduce signal reflectionsis referred to as “terminating” the trace. For example, the inputimpedance of the load may be altered to match the characteristicimpedance of the trace in order to prevent signal reflection. As thetransition time (i.e., the rise or fall time) of the signal decreases,so does the length of trace which must be terminated in order to reducesignal degradation.

[0008] A digital signal alternating between the high and low voltagelevels includes contributions from a fundamental sinusoidal frequency(i.e., a first harmonic) and integer multiples of the first harmonic. Asthe rise and fall times of a digital signal decrease, the magnitudes ofa greater number of the integer multiples of the first harmonic becomesignificant. As a general rule, the frequency content of a digitalsignal extends to a frequency equal to the reciprocal of π times thetransition time (i.e., rise or fall time) of the signal. For example, adigital signal with a 1 nanosecond transition time has a frequencycontent extending up to about 318 MHz.

[0009] All conductors have a certain amount of inductance. The voltageacross the inductance of a conductor is directly proportional to therate of change of current through the conductor. At the high frequenciespresent in conductors carrying digital signals having short transitiontimes, a significant voltage drop occurs across a conductor having evena small inductance. A power supply conductor connects one terminal of anelectrical power supply to a power supply terminal of a device, and aground conductor connects a ground terminal of the power supply to aground terminal of the device. When the device generates a digitalsignal having short transition times, high frequency transient loadcurrents flow in the power supply and ground conductors. Power supplydroop is the term used to describe the decrease in voltage at the powersupply terminal of the device due to the flow of transient load currentthrough the inductance of the power supply conductor. Similarly, groundbounce is the term used to describe the increase in voltage at theground terminal of the device due to the flow of transient load currentthrough the inductance of the ground conductor. When the devicegenerates several digital signals having short transition timessimultaneously, the power supply droop and ground bounce effects areadditive. Sufficient power supply droop and ground bounce can cause thedevice to fail to function correctly.

[0010] Power supply droop is commonly reduced by arranging power supplyconductors to form a crisscross network of intersecting power supplyconductors (i.e., a power supply grid). Such a grid network has a lowerinductance, hence power supply droop is reduced. A continuous powersupply plane may also be provided which has an even lower inductancethan a grid network. Placing a “bypass” capacitor near the power supplyterminal of the device is also used to reduce power supply droop. Thebypass capacitor supplies a substantial amount of the transient loadcurrent, thereby reducing the amount of transient load current flowingthrough the power supply conductor. Ground bounce is reduced by using alow inductance ground conductor grid network, or a continuous groundplane having an even lower amount of inductance. Power supply and groundgrids or planes are commonly placed in close proximity to one another inorder to further reduce the inductances of the grids or planes.

[0011] Electromagnetic interference (EMI) is the term used to describeunwanted interference energies either conducted as currents or radiatedas electromagnetic fields. High frequency components present withincircuits producing digital signals having short transition times may becoupled into nearby electronic systems (e.g., radio and televisioncircuits), disrupting proper operation of these systems. The UnitedStates Federal Communication Commission has established upper limits forthe amounts of EMI products for sale in the United States may generate.

[0012] Signal circuits form current loops which radiate magnetic fieldsin a differential mode. Differential mode EMI is usually reduced byreducing the areas proscribed by the circuits and the magnitudes of thesignal currents. Impedances of power and ground conductors createvoltage drops along the conductors, causing the conductors to radiateelectric fields in a common mode. Common mode EMI is typically reducedby reducing the impedances of the power and ground conductors. Reducingthe impedances of the power and ground conductors thus reduces EMI aswell as power supply droop and ground bounce.

[0013] Within the wide frequency range present within electronic systemswith digital signals having short transition times, the electricalimpedance between any two parallel conductive planes (e.g., adjacentpower and ground planes) may vary widely. The parallel conductive planesmay exhibit multiple electrical resonances, resulting in alternatinghigh and low impedance values. Parallel conductive planes tend toradiate a significant amount of differential mode EMI at theirboundaries (i.e., from their edges). The magnitude of differential modeEMI radiated from the edges of the parallel conductive planes varieswith frequency and is directly proportional to the electrical impedancebetween the planes.

[0014]FIG. 1 is a perspective view of a pair of 10 in.×10 in. squareconductive planes separated by a fiberglass-epoxy composite dielectriclayer. Each conductive plane is made of copper and is 0.0014 in. (1.4mils) thick. The fiberglass-epoxy composite layer separating the planeshas a dielectric constant of 4.0 and is 0.004 in. (4 mils) thick. FIG. 2is a graph of the magnitude of the simulated electrical impedancebetween the pair of rectangular conductive planes of FIG. 1 (log₁₀scale) versus the frequency of a voltage between the planes (log₁₀scale). The graph was created by modeling each square inch of the pairof conductive planes as a matrix of transmission line segments. Theimpedance value was computed by simulating the application of a 1 ampereconstant current between the centers of the rectangular planes, varyingthe frequency of the current, and determining the magnitude of thesteady state voltage between the centers of the rectangular planes.

[0015] As shown in FIG. 2, the magnitude of the electrical impedancebetween the parallel conductive planes of FIG. 1 varies widely atfrequencies above about 20 MHz. The parallel conductive planes exhibitmultiple electrical resonances at frequencies between 100 MHz and 1 GHz,resulting in alternating high and low impedance values. The parallelconductive planes of FIG. 1 tend to radiate substantial amounts of EMIat frequencies where the electrical impedance between the planesanywhere near their peripheries is high.

[0016] It would thus be desirable to provide a power distributionnetwork wherein the electrical impedance between parallel conductiveplanes may be stabilized. Such a network would reduce power supplydroop, ground bounce, and the amount of electromagnetic energy radiatedfrom the edges of the planes. Such impedance stabilization may alsoreduce the need for bypass capacitors.

SUMMARY OF THE INVENTION

[0017] The problems outlined above are in large part solved by aninterconnecting apparatus employing a lossy power distribution networkto reduce power plane resonances. In one embodiment, a printed circuitboard includes a lossy power distribution network formed by a pair ofparallel planar conductors separated by a dielectric layer. The pair ofparallel planar conductors includes a first power supply plane suitablefor use, for example, as a ground plane and a second power supply planesuitable for use, for example, as a power plane (e.g., VCC). Thedielectric layer has a loss tangent value of at least 0.2, andpreferably of at least 0.3. In one embodiment, the dielectric materialbetween the power planes could have a frequency dependent loss tangent,such that a loss tangent value of 0.3 is achieved at and above thelowest resonance frequency of the planes. Due to the relatively largeloss tangent characteristic of the dielectric layer separating the powersupply planes, the electrical impedance characteristics associated withthe power planes may be stabilized, and power plane resonances may bereduced. The printed circuit board may also include one or more signallayers separated from the power planes by respective dielectric layers.The dielectric layers separating the signal layers from the power planesor other signal layers may be associated with much lower loss tangentvalues, such as in the range of 0-0.05. In this manner, high frequencylosses associated with the signal traces may be kept relatively low.

[0018] In another embodiment, power plane resonances are suppressed bydecreasing the thickness of the dielectric material between the powersupply planes to less than 0.5 mils. For example, in one embodiment, theplane separation is preferably reduced to less than 0.2 mils such as,for example, 0.1 mils. In embodiments where the plane separationapproaches 0.1 mils or less, plane resonances may be substantiallysuppressed.

[0019] In various embodiments, the power distribution network of aprinted circuit board or a semiconductor package interconnect mayrequire relatively large currents. For example, it is not uncommon forsystems implemented on printed circuit boards to reach DC currentrequirements of 100 amps or more. Thus, relatively heavy copper or otherconductor layers may be required to handle the large currents. Since astructure that includes very heavy conductive layers on a very thindielectric layer may be associated with manufacturing and handlingproblems, a power distribution network may be provided within a printedcircuit board or package interconnect in which numerous, relatively thinconductive layers are separated by relatively thin dielectric layers.For example, rather than employing a single pair of relatively thick(e.g., 1-2 mils) conductor layers separated by a relatively thick (e.g.,1-2 mils) dielectric layer in the power distribution network of aprinted circuit board, a relatively large number of relatively thin(e.g., 0.05-0.3 mils) dielectric layers with relatively thin (e.g.,0.1-0.2 mils) conductor layers on each side. Alternating conductivelayers in the stack up are connected by vias, every second of themconnecting to one polarity (e.g., ground) and every other connecting tothe other polarity (e.g., VCC). In this manner, the power distributionnetwork may have a relatively low DC resistance to support relativelyhigh currents, while attaining a relatively low high frequency impedancewithout resonances.

[0020] In yet another embodiment, a relatively thin conductive layer isprovided between a pair of relatively thick conductive layers. A firstrelatively thick dielectric layer is provided between one of the thickconductive layers and the thin conductive layer, while a relatively thindielectric layer is provided between the other relatively thick copperconductive layer and the thin conductive layer. A PCB core constructedaccording to this embodiment may be associated with relatively goodmechanical strength and stability and may be capable of supportingrelatively high currents. The structure may further be associated with arelatively low high-frequency impedance without resonances. The thinconductive layer may further be formed in a uniform pattern to createfuses which open if a short occurs through a portion of the thindielectric.

BRIEF DESCRIPTION OF THE DRAWINGS

[0021] Other objects and advantages of the invention will becomeapparent upon reading the following detailed description and uponreference to the accompanying drawings in which:

[0022]FIG. 1 is a perspective view of a pair of 10 in.×10 in. squareconductive planes separated by a fiberglass-epoxy composite dielectriclayer;

[0023]FIG. 2 is a graph of the magnitude of the simulated electricalimpedance |Z|(log₁₀ scale) between the pair of rectangular conductiveplanes of FIG. 1 versus the frequency of a voltage (log₁₀ scale) betweenthe planes;

[0024]FIG. 3 is a perspective view of one embodiment of an electricalinterconnecting apparatus including a set of planar electricalconductors separated by dielectric layers;

[0025] FIGS. 4A-4I are graphs illustrating the magnitude of simulatedelectrical impedance between the parallel conductive planes of FIG. 3versus frequency for different loss tangent values of a dielectriclayer.

[0026] FIGS. 5A-5H are graphs of the magnitude of the simulatedelectrical impedance between the conductive planes of FIG. 3 versusfrequency for different plane separations.

[0027] FIGS. 6A-6E are graphs of the magnitude of the simulatedelectrical impedance between the conductive planes of FIG. 3 versusfrequency for different dielectric and conductor thickness values.

[0028]FIG. 7 is a cross-sectional view of another embodiment of a powerdistribution network employing numerous parallel power and groundplanes.

[0029]FIG. 8 is a cross-sectional view of yet another embodiment of apower distribution network employing a thin dielectric layer to reduceresonance.

[0030]FIG. 9 is a top view of a thin conductive layer where smallregions of the layer are coupled to the rest of the plane with shortnarrow bridges to form a fused structure.

[0031] While the invention is susceptible to various modifications andalternative forms, specific embodiments thereof are shown by way ofexample in the drawings and will herein be described in detail. Itshould be understood, however, that the drawings and detaileddescription thereto are not intended to limit the invention to theparticular form disclosed, but on the contrary, the intention is tocover all modifications, equivalents and alternatives falling within thespirit and scope of the present invention as defined by the appendedclaims.

DETAILED DESCRIPTION OF THE INVENTION

[0032]FIG. 3 is a perspective view of one embodiment of an electricalinterconnecting apparatus 10 including a set of planar electricalconductors illustrated by a first signal plane 14, a ground plane 16, apower plane 18, and a second signal plane 20. Additional layers (e.g.,additional signal layers). may be stacked on top of or beneath theillustrated structure, as desired. Interconnecting apparatus 10 may be,for example, a printed circuit board or an interconnect substrate of asemiconductor device package. Power plane 18 and ground plane 16 arecontinuous across at least a portion of interconnecting apparatus 10.First signal plane 14 and second signal plane 20 are patterned intoelectrically conductive traces to form signal lines that electronicallyconnect to components or contact pads of the interconnecting apparatus.First signal plane 14 and ground plane 16 are separated by a firstdielectric layer 22. Ground plane 16 and power plane 18 are separated bya second dielectric layer 24. Power plane 18 and second signal plane 20are separated by a third dielectric layer 26.

[0033] During use of interconnecting apparatus 10, power plane 18 isconnected to a power terminal of an electrical power supply, and groundplane 16 is connected to a ground terminal of the power supply. Groundplane 16 and power plane 18 are each generally referred to as a powersupply plane. In embodiments where interconnecting apparatus is aprinted circuit board, electronic devices 19 (illustrated in phantom)mounted on the surface of the structure and receive electrical power viaground plane 16 and power plane 18. In embodiments where interconnectingapparatus is an interconnect substrate of a semiconductor package,contact pads 21 (also shown in phantom) associated with signal layer 14may provide electrical connection (including power) to correspondingpads of an integrated circuit contact pads (not shown) on the oppositeside of the apparatus (e.g., formed as a portion of signal layer 20) mayprovide connection to terminals (such as BGA leads) of a device package.

[0034] It is customary to express the dielectric and conductive lossesof signal traces by the following formula:$A^{dB} = {4.35\left( {\frac{R_{s}}{Z_{o}} + {G_{d}Z_{o}}} \right)}$

[0035] where

[0036] A is the attenuation of the matched-terminated trace in dB,

[0037] Rs is the series attenuation at the required frequency,

[0038] Gd is the parallel conductance of the dielectrics at the requiredfrequency,

[0039] Zo is the characteristic impedance of trace.

[0040] Rs is the total series resistance of the conductor at thefrequency of interest, determined by the cross section of conductor. Athigher frequencies, the resistance of conductor increases, becausecurrent tends to flow on the surface, leaving for current conductiononly an effective channel of depth, which is proportional to the inversesquare root of frequency. This effective depth is called the skin depth,and at a first approximation is expressed as:$\delta = \sqrt{\frac{1}{\pi \quad f\quad \sigma \quad \mu}}$

[0041] where

[0042] δ is the skin depth,

[0043] f is the frequency of interest,

[0044] σ is the conductivity of conductor,

[0045] μ is the permeability of mconductor.

[0046] The dielectric losses are usually expressed in terms of losstangent, which is the ratio of conductance and capacitive reactance.From this relationship, the Gd (frequency dependent) conductance issimply GD=loss_tangent*omega*C, where omega is the radian frequency. Theloss tangent is usually a weak function of frequency, and therefore theparallel conductance increases approximately linearly with frequency.

[0047] Though the above expressions are usually valid and are appliedmostly to signal traces under some further restrictive conditions, thesame formulas may be applied to power-distribution planes. This approachis validated by the fact that popular simulation methods use matrices ofone-dimensional transmission lines (traces) to obtain the response oftwo-dimensional power planes.

[0048] From the above loss equation, the required loss tangent toachieve the suppression of resonances can be calculated, for instance,by equating the low-frequency equivalent characteristic impedance of theplanes (sqrt(L/C) and the inverse of the parallel loss conductance (Gd)at the lowest resonance frequency (approximately twice the inverse ofthe propagation delay along one side of the planes). By doing so, arequired loss tangent as 1/PI˜0.3 is obtained. This result isindependent of the size and separation of the planes and of thedielectric constant of the material, and depends only on the ratio ofinverse loss conductance and characteristic impedance (here set to one)at the specified frequency.

[0049] FIGS. 4A-4I are graphs illustrating the magnitude of simulatedelectrical impedance between the parallel conductive planes of FIG. 3versus frequency for different loss tangent values of dielectric layer24. The data depicted in the graphs was obtained assuming 10 inch by 10inch square parallel planes, using 0.7-mil copper conducting planes, alossy dielectric with a dielectric constant of 4, and 2 mils of planeseparation. The impedance profiles are shown with the followingdielectric loss tangent values:

[0050] 0.01 (FIG. 4A)

[0051] 0.03 (FIG. 4B)

[0052] 0.1 (FIG. 4C)

[0053] 0.2 (FIG. 4D)

[0054] 0.3 (FIG. 4E)

[0055] 0.4 (FIG. 4F)

[0056] 0.6 (FIG. 4G)

[0057] 0.8 (FIG. 4H)

[0058] 1.0 (FIG. 4I).

[0059] As illustrated by FIGS. 4A-4I, the ripples in the impedanceprofile gradually decreases as the loss tangent reaches a value of 0.3.There is no significant further change in the impedance profile as theloss tangent increases beyond 0.3.

[0060] In accordance, in one embodiment of the electricalinterconnecting apparatus illustrated by FIG. 3, the dielectric layer 24separating ground plane 16 and power plane 18 is provided with a losstangent of at least 0.2, and preferably of 0.3 or higher for frequenciesat or above the lowest resonance frequency of the planes. In thismanner, power plane resonances may be reduced, and low DC resistance maybe attained.

[0061] It is noted that dielectric materials commonly used in printedcircuit boards have a loss tangent typically of only a few percent(e.g., 0.02) at most. In one embodiment, to keep high frequency signallosses associated with the signal traces of first signal plane 14 andsecond signal plane 20 relatively low, dielectric layers 22 and 26 maybe formed using such a common printed circuit board dielectric materialhaving a relatively low loss tangent of approximately 0.01-0.02 (orgenerally within the range of between 0.00 and 0.05).

[0062] Series conductor losses may also help to suppress resonances. Ingeneral, for signal interconnects, a given series conductor lossprovides higher attenuation at high frequencies if the characteristicimpedance of the interconnect is low. Thus, in one embodiment, to lowerthe characteristic impedance, the separation between ground plane 16 andpower plane 18 is reduced. FIGS. 5A-5H are graphs of the magnitude ofthe simulated electrical impedance between the conductive planes of FIG.3 versus frequency for different plane separations. The profilesdepicted in FIGS. 5A-5H again assume 10 inch by 10 inch square parallelplanes, with 0.7-mil copper and a lossless dielectric having adielectric constant of 4. The impedance profiles are depicted for thefollowing dielectric thicknesses:

[0063] 10 mils (FIG. 5A)

[0064] 4 mils (FIG. 5B)

[0065] 2 mils (FIG. 5C)

[0066] 1 mil (FIG. 5D)

[0067] 0.5 mil (FIG. 5E)

[0068] 0.2 mil (FIG. 5F)

[0069] 0.1 mil (FIG. 5G)

[0070] 0.05 mil (FIG. 5H).

[0071] It is evident from FIGS. 5A-5H that with a plane separationapproaching 0.1 mil and less, the plane resonances are almost totallysuppressed. It is noted that the thinnest dielectric commonly used inmodern printed circuit boards is approximately 2-mils (for example, aZBC2000 core). However, as depicted in FIG. 5C, the impedance profileassociated with a 2-mils dielectric thickness exhibits relatively largeresonances. Accordingly, a printed circuit board having a powerdistribution network as illustrated in FIG. 3 is provided where thethickness of dielectric layer 24 is at most 0.5 mil, and is preferably0.1 mil or less.

[0072] Resonances may also be suppressed by reducing the thickness ofthe conductive layers. FIGS. 6A-6E are graphs of the magnitude of thesimulated electrical impedance between the conductive planes of FIG. 3versus frequency for different dielectric and conductor thicknesses. Theprofiles depicted in FIGS. 6A-6E again assume 10 inch by 10 inch squareparallel planes, with a lossless dielectric having a dielectric constantof 4. The impedance profiles are depicted for the following dielectricand conductor thicknesses:

[0073] 2 mils dielectric, 0.1 mils copper (FIG. 6A)

[0074] 0.2 mils dielectric, 0.2 mils copper (FIG. 6B)

[0075] 0.2 mils dielectric, 0.1 mils copper (FIG. 6C)

[0076] 0.1 mils dielectric, 0.1 mils copper (FIG. 6D)

[0077] 0.05 mils dielectric, 0.05 mils copper (FIG. 6E).

[0078] It is noted that the amount of required copper (or otherconductor) in the planes may be dictated by the DC current requirements.With system currents reaching 100 amps or more, sometimes greater thanone ounce of copper (approximately 1.2 mils) may be required toguarantee good power distribution. The use of very heavy copper orconductor layers on very thin dielectrics, however, may createmanufacturing and handling problems. Thus, as illustrated in FIG. 7, inone embodiment a power distribution network formed by a single groundplane and a single power plane may be replaced by multiple thin (e.g.,0.2 mil or less) conductor layers in parallel each with proportionallyless conductive material in each layer, and with a thin (e.g., 0.2 milor less) dielectric layer between each conductor layer. In FIG. 7, aplurality of alternating ground planes 60 and power planes 62 areseparated by respective thin dielectric layers 64. To ensure resonancesuppression, the thickness of each power supply plane 60 and 62 is nomore than 0.5 mil. For example, in one embodiment, each conductive layer60 and 62 formed by copper has a thickness of 0.1 mil. Additionally,each dielectric layer 64 has a thickness of no more than 0.5 mil. Theground planes 60 are electrically interconnected by a plurality of vias66, and the power planes 62 are electrically interconnected by aplurality of vias 68. It is noted that clearance antipads may be etchedin the conductive layers at respective locations of ground planes 60 andpower plane 62 to prevent shorting. More particularly, to prevent vias66 from providing electrical connections to power planes 62, clearanceantipads may be provided at appropriate locations in each power plane 62to avoid such contact. Similar clearance antipads may be provided withinground planes 60. It is noted that additional vias (not shown) forinterconnecting various signal layers may also be incorporated withinthe structure of FIG. 7, as desired.

[0079]FIG. 7 further illustrates additional dielectric layers 70 whichseparate the power distribution network (formed by the alternatingground and power planes 60 and 62) from signal layers 72. In oneembodiment, the thickness of dielectric layers 70 is at least 1 mil tokeep high frequency signal losses relatively low.

[0080] The power distribution network as illustrated in FIG. 7 mayadvantageously reduce power supply resonances while allowing forrelatively high current capabilities and avoiding manufacturing andhandling problems. For example, consider a situation in which a 2 mildielectric layer with one ounce (1.2 mils) copper planes on each side isreplaced with 11 parallel layers of 0.2 mil dielectric with 0.1 milcopper layers on each side. The original structure (having a 2 mildielectric layer with one ounce [1.2 mils] copper planes on each side)has a thickness of 4.4 mils, and an impedance in the 10-1000 MHz rangeof 8-500 milliohms with resonance peaks and dips. A structure embodiedaccording to FIG. 7 having eleven 0.2 mil thick dielectric layers with0.1 mil copper on each side has approximately the same DC resistance,but its high frequency impedance in the same 10-1000 MHz range may bebelow 3 milliohms without resonances.

[0081] Yet another embodiment is illustrated in FIG. 8. In FIG. 8, apower distribution network includes two relatively thick (e.g., eachbeing at least 1.0 mil thick, such as 1.2 mils) conductive layers 74 and76 to allow for relatively high DC currents. A third, relatively thin(e.g, 0.5 mil or less, such as 0.1 mil), conductive layer 78 is furtherprovided, with a thin (e.g, 0.5 mil or less, such as 0.1 mil) dielectriclayer separating conductive layers 74 and 78, and a relatively thick(e.g, at least 1 mil) dielectric layer 82 separating conductive layers76 and 78. A via 84 electrically interconnects conductive layers 76 and78. Conductive layers 74 and 76 in conjunction with dielectric layer 82provides sufficient copper weight for low resistance and high currentcapability, and also provides for mechanical strength and protects thethin inner layers 78 and 80. The thin conductive layer 78 and dielectriclayer 80 provide for low inductance and loss, efficiently suppressingplane resonances. A via 84 is provided to interconnect conductive layers76 and 78. It is noted that the layered structure of FIG. 8 may beformed before (and independent of) the incorporation of the vias (suchas via 84) which interconnect planes 76 and 78. No antipads internal tothe structure need to be incorporated; only the outer layer (conductor74) needs to be provided with an antipad (or similar isolation) toprovide isolation from the vias. It is also noted that in an alternativeembodiment, an additional thin (e.g., 0.5 mil or less, such as 1 um.)conductor layer and an additional thin (e.g., 0.5 mil or less, such as 1um.) dielectric layer could be incorporated between dielectric layer 82and conductive layer 76 to create a symmetric stack-up structure.

[0082] A further advantage of the structure illustrated in FIG. 8 may beachieved by employing a “fused” construction to deal with local defectsor shorts. For example, as illustrated in FIG. 9, a uniform pattern maybe formed on the thin conductive layer 78, where small regions 90 of thelayer are coupled to the rest of the plane with short narrow bridges 92.Slots 94 which are removed or etched away portions of the conductivelayer separate regions 90. If a short occurs due to a failure or defectin the thin dielectric associated with a particular region 90, thenarrow bridges act like a fuse and opens. This allows the remainder ofthe conductive plane 78 the plane may function properly. The shape andsize of the regular pattern may have forms other than that shown in FIG.9. For frequencies up to a few GHz, a slot dimension of approximately100 mils long and 5 mils wide with a 5 mil gap between adjacent slotsmay be sufficient. Embodiments employing such a fused structure mayrequire dielectric materials which will not carbonize or createconductive particles upon arching.

[0083] While the present invention has been described with reference toparticular embodiments, it will be understood that the embodiments areillustrative and that the invention scope is not so limited. Anyvariations, modifications additions and improvements to the embodimentsdescribed are possible. These variations, modifications, additions andimprovements may fall within the scope of the invention as detailedwithin the following claims.

What is claimed is:
 1. A printed circuit board comprising: a firstconductive layer forming a first power supply plane; a second conductivelayer forming a second power supply plane; a first dielectric layerseparating the first and second conductive layers; a third conductivelayer; and a second dielectric layer separating said second and thirdconductive layers, wherein said first dielectric layer has a thicknessof no more than 0.5 mils.
 2. The printed circuit board as recited inclaim 1 wherein the second dielectric layer has a thickness of at least1 mil.
 3. The printed circuit board as recited in claim 1 wherein saidfirst dielectric layer has a thickness of no more than 0.1 mil.
 4. Theprinted circuit board as recited in claim 1 further comprising a viathat electrically interconnects said second conductive layer and saidthird conductive layer.
 5. The printed circuit board as recited in claim1 wherein the second conductive layer has a thickness of no more than0.5 mils.
 6. The printed circuit board as recited in claim 1 wherein thesecond conductive layer has a thickness of no more than 0.2 mil.
 7. Theprinted circuit board as recited in claim 4 wherein the seconddielectric layer has a thickness of 1 mil or greater.
 8. A printedcircuit board comprising: a first conductive layer forming a first powersupply plane; a second conductive layer forming a second power supplyplane; a first dielectric layer adjacent to and separating the first andsecond conductive layers; a second dielectric layer adjunct to saidsecond conductive layer; a third conductive layer forming a third powersupply plane, wherein the third conductive layer is adjacent to saidsecond dielectric layer; a fourth conductive layer forming a fourthpower supply plane; a third dielectric layer adjacent to and separatingthe third and fourth conductive layers.
 9. The printed circuit board asrecited in claim 3 wherein the first dielectric layer has a thickness ofno more than 0.5 mils.
 10. The printed circuit board as recited in claim9 wherein the first conductive layer has a thickness of no more than 0.5mils.
 11. The printed circuit board as recited in claim 10 wherein thesecond conductive layer, the second dielectric layer, the thirdconductive layer, the fourth conductive layer, and the third dielectriclayer each have a thickness of no more than 0.5 mils.
 12. The printedcircuit board as recited in claim 8 wherein the first conductive layer,the second conductive layer, the third conductive layer, the fourthconductive layer, the first dielectric layer, the second dielectriclayer, and the third dielectric layer each have a thickness of no morethan 0.2 mils.
 13. The printed circuit board as recited in claim 8further comprising a via which electrically interconnects the firstconductive layer and the third conductive layer.
 14. The printed circuitboard as recited in claim 14 further comprising a second via whichelectrically interconnects the second conductive layer and the fourthconductive layer.
 15. A printed circuit board comprising: a firstconductive layer forming a first power supply plane; a second conductivelayer forming a second power supply plane; and a dielectric layerseparating said first and second conductive layers, wherein saiddielectric layer has a loss tangent value of at least 0.2.
 16. Anelectrical interconnecting apparatus comprising: a first conductivelayer forming a first power supply plane; a second conductive layerforming a second power supply plane; a first dielectric layer separatingthe first and second conductive layers; a third conductive layer; and asecond dielectric layer separating said second and third conductivelayers, wherein said first dielectric layer has a thickness of no morethan 0.5 mils.
 17. An electrical interconnecting apparatus as recited inclaim 16 wherein the second dielectric layer has a thickness of at least1 mil.
 18. An electrical interconnecting apparatus as recited in claim16 wherein said first dielectric layer has a thickness of no more than0.1 mil.
 19. The electrical interconnecting apparatus as recited inclaim 16, wherein the electrical interconnecting apparatus forms asubstrate within an integrated circuit package.
 20. An electricalinterconnecting apparatus comprising: a first conductive layer forming afirst power supply plane; a second conductive layer forming a secondpower supply plane; a first dielectric layer adjacent to and separatingthe first and second conductive layers; a second dielectric layeradjunct to said second conductive layer; a third conductive layerforming a third power supply plane, wherein the third conductive layeris adjacent to said second dielectric layer; a fourth conductive layerforming a fourth power supply plane; a third dielectric layer adjacentto and separating the third and fourth conductive layers.
 21. Anelectrical interconnecting apparatus as recited in claim 20 wherein thefirst conductive layer, the second conductive layer, the firstdielectric layer, the second dielectric layer, the third conductivelayer, the fourth conductive layer, and the third dielectric layer eachhave a thickness of no more than 0.5 mils.
 22. The electricalinterconnecting apparatus as recited in claim 20, wherein the electricalinterconnecting apparatus forms a substrate within an integrated circuitpackage.